Gate driving device

ABSTRACT

The present invention copes with fluctuations in a power supply voltage when a capacitor for coping with fluctuations in the power supply voltage has been omitted and also cases in which the power supply voltage is constantly low, thereby ensuring driving of an active element. A gate driving device of an IGBT includes: a first switch portion which turns on the IGBT; a second switch portion which turns off the IGBT; a current control portion which controls the outflow of charge on the gate to a ground line such that current is constant; a first protection circuit which suppresses outflow of gate current to the power supply line; and a second protection circuit which detects a prescribed fluctuation in an internal power supply voltage Vdc, and interrupts the connection between the current control portion and the ground line.

TECHNICAL FIELD

This invention relates to a gate driving device which drives the gate ofan active element with a large input capacitance, such as an IGBT(Insulated Gate Bipolar Transistor), a power MOSFET (Metal OxideSemiconductor Field Effect Transistor), or similar.

BACKGROUND ART

Gate driving devices of this type have been proposed having variousconfigurations, such as for example those described in Japanese PatentApplication Laid-open No. 2008-291728 and Japanese Patent ApplicationLaid-open No. 2010-288444.

Japanese Patent Application Laid-open No. 2008-291728 discloses a devicewhich uses an IGBT to control the current flowing in primary-sidewindings to control ignition by a plug connected to the coilsecondary-side windings, and achieves both reduced turn-on voltage forlow battery voltages and a sufficient conduction start time.

Japanese Patent Application Laid-open No. 2010-288444 discloses a gatedriving device in which, as shown in FIG. 11, the load is taken to be aninductance L, and the gate of an IGBT or other active element having acurrent sensing function is driven by a control IC 4.

In the case of this gate driving device, the inductance L as the loadand an IGBT 3 are connected in series between a power supply line 1 towhich the power supply voltage Vbatt of a battery as an external powersupply and a ground line 2 connected to ground gnd.

Further, the control IC 4, and a current-limiting resistor RB for when avoltage equal to or greater than the clamp voltage of the IC 4 isapplied across A-B, are connected in parallel with the inductance L andIGBT 3. A current sense voltage Vsns output from a current sensingterminal s of the IGBT 3 is input to this control IC 4. The gate voltageVg output from the control IC 4 is supplied to the gate of the IGBT 3.

Further, a noise elimination capacitor C1 which eliminateshigh-frequency noise from the inductance L is connected in parallel withthe series circuit of the current-limiting resistor RB and control IC 4.A high-pass capacitor C2 is connected to the power supply line 1 and theground line 2 in parallel with the control IC 4. L1 and L2 are thewiring inductances of the power supply line 1 and ground line 2.

FIG. 12 shows a specific configuration of a portion related to drivingcontrol of the IGBT 3 by the control IC 4 shown in FIG. 11.

As shown in FIG. 12, in the control IC 4, a constant-current source 5,P-type MOS transistor M1, and N-type MOS transistor M3 are connected inseries between a power supply line 11 to which an internal power supplyvoltage Vdc0 is applied and a ground line 12 connected to ground gnd. AP-type MOS transistor for current control M2 is connected in parallelwith the MOS transistor M3. Parasitic body diodes D1 to D3 are connectedin parallel with the MOS transistors M1 to M3. The op-amp 6 andresistors R1 and R2 form an error amplifier, to control the gate voltageof the MOS transistor M2.

Using a control IC 4 configured in this way, the MOS transistors M1 andM3 are controlled to turn on/off by switch signals SWp, SWn synchronizedwith a control signal Sin input to the control IC 4 shown in FIG. 11, tocontrol charging/discharging of the IGBT 3. That is, when the MOStransistor M1 is turned on the IGBT 3 is charged, and when the MOStransistor M1 is turned off the IGBT 3 is discharged.

The op-amp 6 controls the gate voltage of the MOS transistor M2 suchthat the current sense voltage Vsns obtained by using a sense resistorto convert the sense current input from the current sense terminal S ofthe IGBT 3 into a voltage is equal to the reference voltage Vref; bythis means, the gate voltage Vg of the IGBT 3 is controlled to controlthe collector current Ic of the IGBT 3.

If the amplitude of the battery voltage Vbatt fluctuates as a result ofbattery ripple, the voltage across points A-B in FIG. 11 dropsmomentarily at the time the battery voltage Vbatt falls due to theresonance circuit formed by the wiring inductances L1 and L2 and thecapacitor C1. The momentary drop in voltage gradually increases withincreasing collector current Ic of the IGBT 3, and momentarily fallsbelow the minimum operating power supply voltage of the control IC 4.

However, a bypass capacitor C2 is connected in parallel with the controlIC 4, and a low-pass filter is formed by this bypass capacitor C2 andthe current-limiting resistor RB. Through this low-pass filter effect,the voltage across the bypass capacitor C2 repeatedly undergoes gradualincreases and decreases. Consequently the internal power supply voltageVdc of the control IC 4 is held at a substantially constant voltagewhich sufficiently exceeds the minimum operating power supply voltage ofthe control IC 4.

When the bypass capacitor C2 is eliminated with the object of reducingthe number of components, the low-pass filter effect can no longer beproduced. Hence the voltage across points C-B of the control IC 4 shownin FIG. 11 assumes the same waveform as the voltage across points A-B.Consequently a momentary voltage drop occurs in the internal powersupply voltage Vdc, and the gate voltage Vg of the IGBT 3 also undergoesa large momentary voltage drop. As a result a sharp change is impartedto the current Ic flowing in the IGBT 3, and an induced voltageproportional to the current change occurs in the inductance L which isthe load.

There are two reasons for the momentary large voltage drop in the gatevoltage Vg.

The first reason is that the relation between the internal power supplyvoltage Vdc of the control IC 4 and the gate voltage Vg temporarilychanges to Vdc<Vg, and gate charge accumulated on the gate of the IGBT 3flows out to the power supply line 11 via the parasitic body diode D1 ofthe MOS transistor M1. The second reason is that, if there is a sharpvoltage drop during current-limiting control, gate charge which hasaccumulated on the gate of the IGBT 3 via the MOS transistor M2 flowsout to the ground line 12.

Thus in order to eliminate the bypass capacitor C2, the above-describedproblems must be resolved; as a means of improvement, devices such asthe control IC 4 a shown in FIG. 13 are known (see Japanese PatentApplication Laid-open No. 2010-288444).

As shown in FIG. 13, this control IC 4 a has the basic configuration ofthe control IC 4 shown in FIG. 12, but further adds the followingconstituent elements.

That is, a parallel circuit inserted into the internal power supply line11 and comprising a resistor R3 and a diode D11, and a voltage dropsuppression circuit 80, are added. Further, a diode D12 connected inparallel with the series circuit of the constant-current source 5 andMOS transistor M1, and a resistor R4 inserted between the outputterminal of the op-amp 6 and the gate of the MOS transistor M2, areadded.

The voltage drop suppression circuit 80 comprises a low voltagedetection circuit 8 which detects momentary voltage drops in theinternal power supply voltage Vdc0, and a MOS transistor M4 whichimmediately turns off when the low voltage detection circuit 8 detects avoltage drop. A parasitic body diode D4 is connected in parallel withthe MOS transistor M4.

In the control IC 4 a configured in this way, when the amplitude of thebattery voltage Vbatt fluctuates due to battery ripple, as a resultmomentary overshoot occurs in the internal power supply voltage Vdc0 ofthe control IC 4 a at the time the voltage falls. This overshoot of theinternal power supply voltage Vdc0 increases with increasing collectorcurrent Ic of the IGBT 3, and finally falls below the minimum operatingvoltage of the control IC 4.

That is, due to resonance between the wiring inductances L1 and L2 andthe noise elimination capacitor C1, the terminal voltage Vab calls.Under this influence a state ensues in which the internal power supplyvoltage Vdc0 falls, and momentarily falls below the minimum operatingvoltage of the control IC 4 a.

However, the low voltage detection circuit 8 detects such a momentaryvoltage drop of the internal power supply voltage Vdc0, and immediatelyturns off the MOS transistor M4 based on this detection. Hencedischarging of charge accumulated on the gate of the IGBT 3 to theground line 12 via the MOS transistors M2 and M4 can be reliablyprevented. As a result, the IGBT 3 can continue to operate in the onstate.

On the other hand, when the internal power supply voltage Vdc0 is lowerthan the gate voltage Vg of the IGBT 3 due to a momentary drop in theinternal power supply voltage Vdc0, charge which has accumulated on thegate of the IGBT 3 flows out to the power supply line 11 via theparasitic body diode D1 of the MOS transistor M1 (or the diode D12).

However, the parallel circuit of the diode D11 and the resistor R3 isinserted into the power supply line 11. Hence the diode D11 prevents theoutflow of gate charge to the internal power supply circuit (not shown)connected to the power supply line 11. Further, the resistor R3 forms alow-pass filter with the capacitance of the gate of the IGBT 3, and somomentary movement of gate charge toward the internal power supplycircuit is prevented, and in addition the minimum limiting current atwhich operation of the circuit connected to the internal power supply ispossible is supplied.

As explained above, when a momentary voltage drop occurs in the internalpower supply voltage Vdc0 within the control IC 4 a shown in FIG. 13,outflow of gate charge from the IGBT 3 to the internal power supplycircuit and outflow to the ground line 12 can both be reliablysuppressed. Hence the IGBT 3 can hold the gate charge and suppress dropsin the gate voltage Vg, and the internal power supply voltage Vdc on thedownstream side of the parallel circuit of the diode D11 and resistor R3in the control IC 4 a can be held at a voltage slightly reduced from thevoltage immediately before the drop in the internal power supply voltageVdc.

Moreover, despite large fluctuations in the internal power supplyvoltage Vdc0, extremely small fluctuations in the internal power supplyvoltage Vdc can be suppressed and the internal power supply can bestabilized. Hence another purpose can be served of temporarilyaugmenting (supplying) the voltage for another circuit which uses theinternal power supply voltage Vdc as a power supply.

However, in the control IC 4 a shown in FIG. 13, when for example thepower supply voltage is constantly at low voltage, it is no longerpossible to ignore voltage drops across the point of supply of theinternal power supply voltage Vdc0 of the power supply line 11 and thegate of the IGBT 3, and there is the possibility that the gate voltageVg of the IGBT 3 may drop and the IGBT 3 can no longer be drivenadequately.

On the other hand, a control IC 4 b such as that shown in FIG. 14 hasbeen proposed.

This control IC 4 b adds a regulator circuit 7 to the configuration ofthe control IC 4 a shown in FIG. 13. The regulator circuit 7 stabilizesthe voltage Vbin across points C-B in FIG. 11, and outputs a stabilizedvoltage Vreg.

FIG. 15 shows a specific example of the low voltage detection circuit 8shown in FIG. 14 (see Japanese Patent Application Laid-open No.2010-288444).

As shown in FIG. 15, this low voltage detection (undervoltage detector)circuit 8 comprises a self-biased type comparator 81, connected betweenthe internal power supply line 11 to which the internal power supplyvoltage Vdc is supplied and the ground line 12 connected to ground gnd.

The non-inverting input of this comparator 81 is connected to theconnection point between a resistor R11 and an N-type MOS transistorM11, which are connected in series between the internal power supplyline 11 and the ground line 12. The inverting input of the comparator 81is connected to the connection point between a diode D31 and a resistorR13, which are connected in series between the line 13 to which theoutput voltage Vreg of the regulator circuit 7 is applied and the groundline 12.

A parallel circuit of an inverter 82, a resistor R14 and a diode D32 thecathode of which is on the side of the comparator 81 is inserted betweenthe output side of the comparator 81 and the N-type MOS transistor M4,and a gate signal is output from this parallel circuit to the gate ofthe MOS transistor M4. C10 is the gate-emitter capacitance of the MOStransistor M4.

Next, an example of operation of the low voltage detection circuit 8shown in FIG. 15 is explained, referring to FIG. 14 to FIG. 16.

As shown in FIG. 16(A), the voltage Vbin across points C-B in FIG. 11 israised (increased) from the minimum value (0 V) to the maximum value,and upon reaching the maximum value, falls (decreases) to the minimumvalue.

This change in the voltage Vbin is accompanied by a rise in the outputvoltage Vreg of the regulator circuit 7 shown in FIG. 14 and theinternal power supply voltage Vdc, while maintaining the relationVreg>Vdc, as shown in FIG. 16(A); each reaches a constant value andmaintains the constant value. Thereafter, each falls from the respectiveconstant value to the minimum value.

This change is accompanied by a change in the input voltage V+ of thenon-inverting input terminal (+) and the input voltage V− of theinverting input terminal (−) of the comparator 81 shown in FIG. 15, asshown in FIG. 16(B), and the input voltage V− is always higher than theinput voltage V+. This is because the rise in the output voltage Vreg isearlier than that of the internal power supply voltage Vdc, and moreoverthe forward-direction voltage of the diode D31 is lower than thethreshold voltage of the MOS transistor M11.

As a result the output CMPout of the comparator 81 goes to L level (lowlevel), as shown in FIG. 16(C). In FIG. 16(C), where there is a minutevoltage, an indeterminate state occurs at the rising or falling of thepower supply.

The output of the comparator 81 is logically inverted by the inverter82, and so the output of the inverter 82 is H level (high level). Hencethe output voltage OUTB of the low voltage detection circuit 8 shown inFIG. 15 is always H level, as shown in FIG. 16(D), and is applied to thegate of the MOS transistor M4 shown in FIG. 14.

As a result, the drain current Id of the MOS transistor M4 is thecurrent shown in FIG. 16(E). That is, in the state at which the powersupply voltage is low (low power supply voltage state), the VREFpotential, which ordinarily should be constant, falls, and together withthis the charge which should have charged the gate of the IGBT 3 flowsout to the ground line 12 via the MOS transistor M4 due to the operationof the op-amp 6, so that the voltage at the gate of the IGBT 3 falls. Asa result, when in the low power supply voltage state, the current Icflowing in the IGBT 3 is limited to a low current range.

DISCLOSURE OF THE INVENTION

The present invention focuses on the above-described problems, and hasas an object the provision of a gate driving device which can addressfluctuations in a power supply voltage and secure driving of an activeelement even when a capacitor necessary to address fluctuations in thepower supply voltage is omitted, and moreover which can secure drivingof the active element even when the power supply voltage is constantlyat a low power supply voltage.

In order to attain the above-described object, one mode of the inventionis a gate driving device which drives a gate of an active element with alarge input capacitance, this device including: a first switch portionwhich is provided between a high potential side first power supply lineand a gate of the active element and which turns on the active element;a second switch portion which is connected between the gate of theactive element and a low potential side second power supply line andwhich turns off the active element; a current control portion which isprovided in parallel with the second switch portion and which controlsoutflow of charge on a gate of the active element to the second powersupply line such that a current that flows in the active element isconstant; a first protection circuit which is provided between the firstswitch portion and the gate of the active element and which suppressesoutflow of charge on the gate of the active element to the first powersupply line; and a second protection circuit which is provided betweenthe current control portion and the second power supply line and whichdetects a prescribed fluctuation in an applied voltage between the firstpower supply line and the second power supply line, and when thefluctuation is detected, interrupts the connection between the currentcontrol portion and the second power supply line.

In another mode of this invention, the first switch portion includes afirst transistor, and the first transistor operates as aconstant-current source when the active element is turned on and haltsthe operation as a constant-current source when the active element isturned off.

Further, in another mode of this invention, the first switch portionincludes a second transistor which together with the first transistorforms a current mirror circuit, and a third transistor which isconnected in series with the second transistor and which turns on andoff according to whether the active element is on or off.

Further, in another mode of this invention, the second protectioncircuit comprises a diode which prevents outflow of charge on the gateof the active element to the first power supply line, and a resistorwhich is connected in parallel with the diode and which forms a low-passfilter with the gate capacitance of the active element.

Another mode of this invention is a gate driving device which drives agate of an active element with a large input capacitance, this deviceincluding: a first switch portion which is provided between a highpotential side first power supply line and the gate of the activeelement and which turns on the active element; a second switch portionwhich is connected between the gate of the active element and a lowpotential side second power supply line and which turns off the activeelement; a current control portion which is provided in parallel withthe second switch portion and which controls outflow of charge on thegate of the active element to the second power supply line such that acurrent that flows in the active element is constant; a first protectioncircuit which is provided between an external power supply and the firstpower supply line and which suppresses outflow of charge on the gate ofthe active element to the external power supply; and a second protectioncircuit which is provided between the current control portion and thesecond power supply line and which, upon detecting a momentary drop in avoltage of the external power supply or upon detecting that a voltage ofthe external power supply is in a low power supply voltage state,interrupts connection between the current control portion and the secondpower supply line.

Thus in one mode of the invention, a first protection circuit and asecond protection circuit are provided, so that a bypass capacitor canbe omitted, and even when the power supply voltage fluctuates,fluctuation of the gate voltage of the active element can be suppressedto the extent possible, and adequate driving of the gate of the activeelement can be secured.

Further, in one mode of the invention, a first switch portion and afirst protection circuit only are provided between the first powersupply line and the active element gate, so that compared with the priorart, voltage drops between the first power supply line and the gate ofthe active element can be reduced. Consequently driving of the activeelement can be secured even when the power supply voltage is constantlya low power supply voltage.

Further, in one mode of the invention, a second protection circuit, upondetecting that the voltage of an external power supply is in a low powersupply voltage state, interrupts the connection between the currentcontrol portion and the second power supply line. Hence in low powersupply voltage operation the active element can be driven withoutlimiting the current flowing therein, without drops in the voltage inputto the active element.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram showing a first embodiment of a gate drivingdevice of the invention;

FIG. 2 is a circuit diagram of the control IC shown in FIG. 1;

FIG. 3 is a waveform diagram of different portions to explain operationof the control IC shown in FIG. 2;

FIG. 4 is a circuit diagram showing the configuration of a modifiedexample of the control IC shown in FIG. 2;

FIG. 5 is a circuit diagram of the control IC of a second embodiment ofa gate driving device of the invention;

FIG. 6 is a circuit diagram showing the specific configuration of thelow voltage detection circuit shown in FIG. 5;

FIG. 7 is a waveform diagram of different portions to explain operationcorresponding to direct-current voltage changes in the low voltagedetection circuit shown in FIG. 6;

FIG. 8 is a waveform diagram of different portions to explain operationcorresponding to transient voltage changes in the low voltage detectioncircuit shown in FIG. 6;

FIG. 9 is a waveform diagram of different portions to explain operationwhen the power supply voltage in the second embodiment is at lowvoltage;

FIG. 10 is a waveform diagram of different portions to explain operationwhen a momentary voltage drop occurs in the power supply voltage in thesecond embodiment;

FIG. 11 is a circuit diagram showing the configuration of a device ofthe prior art;

FIG. 12 is a circuit diagram of the control IC shown in FIG. 11;

FIG. 13 is a circuit diagram of a control IC of the prior art, as animprovement over the control IC shown in FIG. 12;

FIG. 14 is a circuit diagram of a control IC, as an improvement over thecontrol IC shown in FIG. 13;

FIG. 15 is a circuit diagram showing a specific configuration of the lowvoltage detection circuit shown in FIG. 14; and

FIG. 16 is a waveform diagram of different portions to explain operationcorresponding to direct-current voltage changes in the low voltagedetection circuit shown in FIG. 15.

BEST MODE FOR CARRYING OUT THE INVENTION

Below, embodiments of the invention are explained based on the drawings.

Configuration of a First Embodiment

FIG. 1 is a circuit diagram showing a first embodiment of a gate drivingdevice of the invention.

As shown in FIG. 1, the first embodiment of a gate driving device drivesthe gate of an active element with a large input capacitance such as anIGBT 3, and substitutes the control IC 4 c for the control IC 4 in FIG.11, as well as eliminating the bypass capacitor C2 in FIG. 11.

That is, in the first embodiment, an inductance L as the load and anIGBT 3 are connected in series between the power supply line 1 to whichthe battery power supply voltage Vbatt is applied as an external powersupply, and the ground line 2 connected to ground gnd.

The control IC 4 c and a current-limiting resistor RB for when a voltageequal to or greater than the clamp voltage of the IC 4 c is appliedacross the points A-B are connected in parallel with the inductance Land the IGBT 3. A current sense voltage Vsns output from the currentsense terminal s of the IGBT 3 is input to the control IC 4 c. The gatevoltage Vg output from the control IC 4 c is supplied to the gate of theIGBT 3.

Further, a noise elimination capacitor C1 which eliminateshigh-frequency noise from the inductance L is connected in parallel withthe series circuit of the current-limiting resistor RB and control IC 4c. L1 and L2 are the wiring inductances of the power supply line 1 andthe ground line 2.

(Configuration of the Control IC)

Next, a specific configuration of the gate control portion of thecontrol IC 4 c shown in FIG. 1 is explained, referring to FIG. 2.

In the control IC 4 c, a gate control portion is connected as shown inFIG. 2 between the internal power supply line 11 connected to aregulator circuit (not shown) which generates an internal power supplyvoltage Vdc (=Vreg) based on the battery power supply voltage Vbattinput via the current-limiting resistor RB shown in FIG. 1, and theground line 12 connected to ground gnd.

As shown in FIG. 2, this gate control portion comprises a first switchportion 20, first protection circuit 30, second switch portion 40,current control portion 50, and second protection circuit 60. Thesecomponents are provided between the power supply line 11 which is thepower supply line on the high potential side, and the ground line 12which is the power supply line on the low potential side.

Specifically, the first switch portion 20, first protection circuit 30and second switch portion 40 are connected in series between the powersupply line 11 and the ground line 12. Further, the connection portioncommon to the first protection circuit 30 and the second switch portion40 is connected to the output terminal 70, and this output terminal 70is connected to the gate of the IGBT 3. The current control portion 50and second protection circuit 60 are connected in series parallel to thesecond switch portion 40.

The first switch portion 20 is provided between the power supply line 11and the first protection circuit 30, and comprises a switching elementSW which is controlled to turn on and off by a switch signal Swp. Thisswitching element SW, when turned on, charges the gate of the IGBT 3.

The first protection circuit 30 suppresses the outflow of charge on thegate of the IGBT 3 to the power supply line 11, and forms a low-passfilter with the gate capacitance of the IGBT 3, and comprises a parallelcircuit of a diode D21 and resistor R5 connected in parallel. One end ofthis parallel circuit is connected to the switching element SW and theother end is connected to the output terminal 70 connected to the gateof the IGBT 3.

The second switch portion 40 comprises an N-type MOS transistor M3; thisMOS transistor M3 includes a parasitic body diode D3 connected inparallel. The parallel circuit of the MOS transistor M3 and diode D3 isconnected between the output terminal 70 and the ground line 12. Whenthe MOS transistor M3 is turned on, the charge on the gate of the IGBT 3is discharged. Here, a diode D22 which provides a clamp function duringconduction (turn-on) of the switching element SW is connected inparallel with the second switch portion 40.

The current control portion 50 is provided in parallel with the secondswitch portion 40, and controls outflow of charge on the gate of theIGBT 3 to the ground line 12 such that the collector current flowing inthe IGBT 3 is constant.

To this end, the current control portion 50 comprises a MOS transistorM3 connected between the output terminal 70 and the MOS transistor M4 ofthe second protection circuit 60, and an error amplifier 51 whichcontrols the gate voltage of this MOS transistor M2. The error amplifier51 comprises an op-amp 6 and resistors R1, R2 and R4. The erroramplifier 51 generates a voltage according to the difference between thecurrent sense voltage Vsns input from the current sense terminal s ofthe IGBT 3 and the reference voltage Vref, and outputs this voltage tothe MOS transistor M2.

The second protection circuit 60 is provided between the current controlportion 50 and the ground line 12, and detects prescribed fluctuationsin the internal power supply voltage Vdc, and upon detecting such afluctuation, interrupts the connection between the current controlportion 50 and the ground line 12.

To this end, the second protection circuit 60 comprises a low voltagedetection circuit 61 which detects momentary voltage drops in theinternal power supply voltage Vdc, and a MOS transistor M4 whichimmediately turns off when this low voltage detection circuit 61 detectsa voltage drop. A parasitic body diode D4 is connected in parallel withthe MOS transistor M4. This parallel circuit is connected between theMOS transistor M2 of the current control portion 50 and the ground line12.

(Operation of the Control IC)

Next, an example of operation of the control IC 4 c is explained,referring to FIG. 1 to FIG. 3.

When, at time t0 in FIG. 3, the battery voltage Vbatt shown in FIG. 1 isa prescribed constant voltage and is normal, as shown in FIG. 3( a), thevoltage across points A-B and the voltage across points C-B shown inFIG. 1 are as shown in FIG. 3( b) and FIG. 3( c). Hence the internalpower supply voltage Vdc applied to the internal power supply line 11 inFIG. 2 is as shown in FIG. 3( e), and is substantially equal to thebattery voltage Vbatt.

At this time, when the logical value of the control signal Sin input tothe gate of the control IC 4 c is “Low”, as shown in FIG. 3( d), thelogical values of the switch signal SWp which turns the switchingelement SW on and off and of the switch signal SWn input to the MOStransistor M3 are “High”, as shown in FIG. 3( f). Hence the switchingelement SW enters the off state, and the MOS transistor M3 enters the onstate.

Consequently the gate voltage Vg of the IGBT 3 goes to ground level, asshown in FIG. 3( h), so that the IGBT 3 enters the off state and thecurrent Ic flowing in the IGBT 3 transitions to “0”, as shown in FIG. 3(i). As a result, the current sense voltage Vsns output from the currentsense terminal s of the IGBT 3 also transitions to “0”, as shown in FIG.3( g).

Thereafter, at time t1 in FIG. 3, when the logical value of the controlsignal Sin changes from “Low” to “High” as shown in FIG. 3( d), thelogical values of the switch signals SWp and SWn are inverted from“High” to “Low” as shown in FIG. 3( f). In accordance with this, theswitching element SW enters the on state and the MOS transistor M3 isturned off, so that the gate voltage Vg of the IGBT 3 becomes a positivevoltage substantially coinciding with the internal power supply voltageVdc, as shown in FIG. 3( h).

Consequently the IGBT 3 enters the on state, and the current Ic flowingin the IGBT 3 gradually increases, as shown in FIG. 3( i). In accordancewith this, the current sense voltage Vsns generated based on the sensecurrent output from the current sense terminal s of the IGBT 3 graduallyincreases, as shown in FIG. 3( g). Thereafter the current controlportion 50 performs current limiting processing such that the sensevoltage Vsns substantially coincides with the reference voltage Vref.

Suppose that at time t2 in FIG. 3, a voltage fluctuation has occurred inwhich the battery voltage Vbatt undergoes a voltage drop repeatedly witha prescribed period, as shown in FIG. 3( a).

In this case, in accordance with the voltage fluctuations in the batteryvoltage Vbatt, the internal power supply voltage Vdc in the control IC 4c undergoes momentary overshoot at the point of the falling edge of thevoltage fluctuation, as shown in FIG. 3( e), and this overshootincreases with increasing collector current Ic, finally falling belowthe minimum operating voltage of the control IC 4 c.

That is, due to the resonance of the wiring inductances L1, L2 and thenoise elimination capacitor C1, shown in FIG. 1, the voltage across thepoints A-B in FIG. 1 falls (see FIG. 3( b)). Under this influence theinternal power supply voltage Vdc falls and momentarily becomes lessthan the minimum operating voltage of the control IC 4 c, as shown inFIG. 3( e).

However, when fluctuations in the internal power supply voltage Vdcbegin, the low voltage detection circuit 61 detects a momentary voltagedrop in the internal power supply voltage Vdc, and as a result of thisdetection, the low voltage detection circuit 61 immediately turns offthe MOS transistor M4.

Hence even when the current control portion 50 executes conductioncontrol of the MOS transistor M2 such that the current sense voltageVsns is made to coincide with the reference voltage Vref, the MOStransistor M4 continues in the off state. As a result, discharging ofcharge accumulated on the gate of the IGBT 3 to the ground line 12through the MOS transistor M2 can be reliably prevented. By this means,the IGBT 3 can be made to continue in the on state.

On the other hand, if the gate voltage Vg of the IGBT 3 becomes higherthan the internal power supply voltage Vdc due to a momentary sharp dropin the internal power supply voltage Vdc, charge accumulated on the gateof the IGBT 3 attempts to flow out to the internal power supply line 11.

However, the first protection circuit 30, in which the resistor R5 anddiode D21 are connected in parallel, is inserted between the outputterminal 70 (gate of the IGBT 3) and the switching element SW. Hence theoutflow of accumulated charge to the internal power supply line 11 isprevented by the diode D21. Further, the gate capacitance of the IGBT 3and the resistor R5 form a low-pass filter, so that momentary movementof the above-described accumulated charge to the internal power supplyline 11 can be prevented, and the minimum current necessary to enableoperation of the circuit connected to the internal power supply isprovided.

In this way, in the control IC 4 c shown in FIG. 2, when a momentaryvoltage drop has occurred in the internal power supply voltage Vdc,outflow of charge on the gate of the IGBT 3 to both the internal powersupply line 11 and to the ground line 12 can be reliably suppressed. Asa result, for the IGBT 3 the gate charge can be held and drops in thegate voltage Vg can be suppressed.

Consequently, the gate voltage Vg of the IGBT 3 can be limited tosmall-amplitude fluctuations without any sharp drops, as shown in FIG.3( h). In accordance with this, the current Ic flowing in the IGBT 3 canbe made to increase stably, as shown in FIG. 3( i). Hence the occurrenceof induced voltages proportional to current changes in the inductance Las the load can be reliably prevented.

Modified Example of the Control IC

Next, the configuration of a modified example of the control IC isexplained, referring to FIG. 4.

As shown in FIG. 4, the control IC 4 d of this modified example is basedon the configuration of the control IC 4 c shown in FIG. 2, butsubstitutes the configuration of the first switch portion 20 shown inFIG. 4 for the switching element SW of FIG. 2.

Hence portions which are the same as constituent elements of the controlIC 4 c shown in FIG. 2 are assigned the same symbols, and explanationsare omitted.

As shown in FIG. 4, the first switch portion 20 of the control IC 4 dcomprises an N-type MOS transistor M5 and P-type transistors M6 and M7.Each of the MOS transistors M5 to M7 is connected in parallel withparasitic body diodes D5 to D7.

In the first switch portion 20, the MOS transistor M7 is made tofunction as a constant-current source, the MOS transistor M5 is turnedon and off by the switch signal SWp, and by means of this on-offoperation, the above-described constant-current source function isturned on and off.

Specifically, the MOS transistor M7 is provided between the internalpower supply line 11 and the first protection circuit 30. Further, theMOS transistor M7 forms a current mirror circuit with the MOS transistorM6. And, the MOS transistor M5 is connected in series with the MOStransistor M6, and the MOS transistor M5 is controlled to turn on andoff by the switch signal SWp.

Here the power supply voltage of the circuit which generates the switchsignal SWp supplied to the gate of the MOS transistor M5 is lower thanthe power supply voltage Vdc of the gate control portion of the controlIC 4 d, and so the MOS transistor M5 level-shifts the switch signal SWp.

By means of this configuration, by arbitrarily setting the transistorsize ratio (mirror ratio) of the MOS transistor M6 and the MOStransistor M7, the current flowing in the MOS transistor M7 can be setto an arbitrary value. And by using the switch signal SWp to turn on theMOS transistor M5, the MOS transistor M7 can be made to function as aconstant-current source.

Further, when using the MOS transistor M7 as a constant-current source,the diode D22 serves to limit the current during clamping.

Advantageous Results of the First Embodiment

As explained above, in the first embodiment a first protection circuit30 and a second protection circuit 60 are provided. Hence in a state inwhich the parallel bypass capacitor is eliminated in the control IC 4 c,even when the power supply voltage input to the control IC 4 cmomentarily falls below the minimum operating voltage, fluctuations inthe gate voltage Vg of the IGBT 3 are suppressed to the extent possible,and adequate driving of the gate of the IGBT 3 can be secured.

Further, in the first embodiment, as shown in FIG. 2, only a switchingelement SW (MOS transistor M7) and first protection circuit 30 areprovided between the internal power supply line 11 and the outputterminal 70, so that compared with the prior art, voltage drops betweenthe internal power supply line 11 and the output terminal 70 can be keptsmall (see FIG. 13). As a result, even when for example the internalpower supply voltage Vdc is constantly at a low power supply voltage,the gate voltage Vg of the IGBT 3 can be secured, and adequate drivingof the IGBT 3 is possible.

Modified Example of the First Embodiment

(1) In the control IC of FIG. 2 and FIG. 4, the internal power supplyvoltage Vdc applied to the internal power supply line 11 was generatedby an internal power supply circuit (not shown) based on a battery powersupply voltage Vbatt.

However, in place of this, the voltage applied to the internal powersupply line 11 may be the voltage across the points C-B in FIG. 1; inthis case also, the above-described advantageous results of the firstembodiment of the invention can be realized.

Configuration of a Second Embodiment

A second embodiment of a gate driving device of the invention is basedon the configuration of the first embodiment shown in FIG. 1, but withthe control IC 4 d shown in FIG. 5 substituted for the control IC 4 cshown in FIG. 1 and FIG. 2.

(Configuration of the Control IC)

Next, a specific configuration of the gate control portion of thecontrol IC 4 d is explained, referring to FIG. 5.

The gate control portion of the control IC 4 d is connected between theinternal power supply line 11 and the ground line 12 connected to groundgnd, as shown in FIG. 5.

That is, as shown in FIG. 5, the gate control portion of the control IC4 d comprises a regulator circuit 7, first protection circuit 30 a,first switch portion 20 a, second switch portion 40, current controlportion 50, and second protection circuit 60 a. These, except for thefirst protection circuit 30 a, are provided between the high potentialside power supply line 11 and the low potential side ground line 12.

Specifically, the first switch portion 20 a and second switch portion 40are connected in series between the power supply line 11 and the groundline 12. The connection portion common to the first switch portion 20 aand the second switch portion 40 is connected to the output terminal 70,and the output terminal 70 is connected to the gate of the IGBT 3. Thecurrent control portion 50 and second protection circuit 60 a areseries-connected in parallel with the second switch portion 40.

The regulator circuit 7 and first protection circuit 30 a are insertedinto the power supply line 11. The regulator circuit 7 takes as inputthe voltage Vbin across points C-B in FIG. 1, and generates and outputsan output voltage Vreg which is stabilized based on the input voltage.This stabilized output voltage Vreg is supplied to the first protectioncircuit 30 a.

The first protection circuit 30 a comprises a parallel circuit of thediode D11 and resistor R3, and prevents or suppresses outflow of chargeon the gate of the IGBT 3 to the regulator circuit 7.

The first switch portion 20 a is provided between the power supply line11 and the output terminal 70, and comprises a P-type MOS transistor M1controlled to turn on and off by the switch signal Swp and a currentsource 5 connected in series with same. When turned on, the MOStransistor M1 charges the gate of the IGBT 3. A parasitic body diode D1and a diode D12 are each connected in parallel with the series circuitof the MOS transistor M1 and the current source 5.

The second switch portion 40 comprises an N-type MOS transistor M3 whichis controlled to turn on and off by the switch signal Swn, and includesa parasitic body diode D3 connected in parallel with the MOS transistorM3. The parallel circuit of the MOS transistor M3 and the diode D3 isconnected between the output terminal 70 and the ground line 12. Whenturned on, the MOS transistor M3 causes discharge of the charge on thegate of the IGBT 3.

The current control portion 50 controls the outflow of charge on thegate of the IGBT 3 to the ground line 12 such that the collector currentflowing in the IGBT 3 is constant.

To this end, the current control portion 50 comprises a MOS transistorM2 connected between the output terminal 70 and the MOS transistor M4 ofthe second protection circuit 60 a, and an error amplifier 51 whichcontrols the gate voltage of the MOS transistor M2. The error amplifier51 comprises an op-amp 6 and resistors R1, R2 and R4. The erroramplifier 51 generates a voltage according to the difference between thecurrent sense voltage Vsns input from the current sense terminal s ofthe IGBT 3 and the reference voltage Vref, and outputs the generatedvoltage to the MOS transistor M2.

The second protection circuit 60 a is provided between the currentcontrol portion 50 and the ground line 12. The second protection circuit60 a detects momentary voltage drops in the voltage Vbin of the externalpower supply, and upon detecting that the power supply voltage is in alow power supply voltage state lower than normal, interrupts theconnection between the current control portion 50 and the ground line12.

To this end, the second protection circuit 60 a comprises a low voltagedetection circuit 62 which detects momentary voltage drops in thevoltage Vbin of the external power supply and a low power supply voltagestate thereof, and a MOS transistor M4 which immediately turns off whenthe low voltage detection circuit 62 detects these. The MOS transistorM4 is connected in parallel to a parasitic body diode D4. This parallelcircuit is connected between the MOS transistor M2 of the currentcontrol portion 50 and the ground line 12.

Next, a specific configuration of the low voltage detection circuit 62shown in FIG. 5 is explained, referring to FIG. 6.

As shown in FIG. 6, the low voltage detection circuit 62 comprises areference voltage generation circuit 621, a voltage detection circuit622, a self-biased type comparator 623, an output circuit 624, and anoutput terminal 625.

The reference voltage generation circuit 621 is connected to theinternal power supply voltage Vdc which is the voltage of the powersupply line 11, and generates a voltage with reference to ground gnd;this voltage is smoothed and output to the non-inverting input terminal(+) of the comparator 623.

The voltage detection circuit 622 takes as input the output voltage Vregof the regulator circuit 7, generates a voltage according to this inputvoltage, and outputs the generated voltage to the inverting inputterminal (−) of the comparator 623.

Here, the output voltage of the reference voltage generation circuit 621(the input voltage V+ of the non-inverting input terminal (+) of thecomparator 623) is set so that, compared with the output voltage of thevoltage detection circuit 622 (the input voltage V− of the non-invertingterminal (−) of the comparator 623), rising is early, and after rising,is low compared with the output voltage of the voltage detection circuit622 (see FIG. 7(B)). Further, when there is a momentary drop in thevoltage Vbin of the external power supply, the drop in the outputvoltage of the reference voltage generation circuit 621 is small, andthe drop in the output voltage of the voltage detection circuit 622 islarge (see FIG. 8(C)).

Specifically, the reference voltage generation circuit 621 comprises avoltage division circuit 6211 which divides the internal power supplyvoltage Vdc, and a low-pass filter 6212 which smoothes and outputs thedivided voltage of the voltage division circuit 6211.

The voltage division circuit 6211 comprises a series circuit of aresistor R11 and a diode-connected transistor M11 connected in series;this series circuit is connected between the power supply line 11 andthe ground line 12. The connection portion common to the voltagedivision circuit 6211 is connected to the input terminal of the low-passfilter 6212. The output terminal of the low-pass filter 6212 isconnected to the non-inverting input terminal (+) of the comparator 623.

The voltage detection circuit 622 comprises a voltage division circuitwhich takes as input the output voltage Vreg of the regulator circuit 7,and divides this input voltage. The voltage division circuit comprises aseries circuit in which a diode D33, resistor R12, and resistor R13 areconnected in series.

In this series circuit, the anode of the diode D33 is connected to thepower supply line 11 and to the line 13 to which the voltage Vreg issupplied, and one end of the resistor R13 is connected to the groundline 12. The connection portion common to the resistor R12 and theresistor R13 is connected to the inverting input terminal (−) of thecomparator 623. And, the portion at which the diode D33 and resistor R12are series connected is connected antiparallel to the diode D31. Thatis, the anode of the diode D31 is connected to one end of the resistorR12, and the cathode is connected to the anode of the diode D33.

Here, the series circuit in which the diode D33, resistor R12 andresistor R13 are connected in series has resistance values and similarset so as to determine the DC level of the input voltage V− of theinverting input terminal (−) of the comparator 623. Further, when thepower supply voltage momentarily drops, the diode D31 can momentarilylower the input voltage V− of the inverting input terminal (−) of thecomparator 623.

Further, the same functions can be realized if the anode of the diodeD33 is connected not to the cathode of the diode D31, but to theinternal voltage Vdc, as indicated by the dashed line in FIG. 6.

The comparator 623 is driven by the internal power supply voltage Vdcapplied to the power supply line 11, compares the output voltage of thereference voltage generation circuit 621 and the output voltage of thevoltage detection circuit 622, generates a signal according to thecomparison result, and outputs the generated signal to the outputcircuit 624.

The output circuit 624 generates a signal to turn the MOS transistor M4on or off based on the output signal of the comparator 623.

To this end, the output circuit 624 comprises an inverter 6241, aresistor R14, a diode D32, and a depression type MOS transistor M12.

The inverter 6241 logically inverts the output signal of the comparator623, and outputs this logically inverted signal. The resistor R14 isconnected between the output side of the inverter 6241 and the outputterminal 625, and the diode D32 is connected in parallel with theresistor R14.

The depression type MOS transistor M12 reliably clamps the voltage ofthe output terminal 625 at ground gnd in the low power supply voltagerange equal to or less than the threshold voltage of the inverter 6241,and connects the output terminal 625 and the ground line 12.

In FIG. 6, C10 is the capacitance between the output terminal 625 andground gnd, and forms a filter with the resistor R14 during charging,but during discharging, serves to remove the charge immediately via thediode D32.

Next, in the example of operation of the low voltage detection circuit62 shown in FIG. 6, operation in a case in which the power supplyvoltage changes to DC voltage is explained with reference to FIG. 5 toFIG. 7.

As shown in FIG. 7(A), the voltage Vbin across points C-B in FIG. 1 israised (increased) from the minimum value (0 V) to the maximum value,and after reaching the maximum value is lowered (reduced) to the minimumvalue.

Such changes in the voltage Vbin are accompanied by rises in the outputvoltage Vreg of the regulator circuit 7 shown in FIG. 5 and the internalpower supply voltage Vdc, while maintaining the relationship Vreg>Vdc,as shown in FIG. 7(A); each then becomes a constant value and ismaintained as a constant value. Thereafter, they fall from theirconstant values to the minimum values.

These changes are accompanied by the changes shown in FIG. 7(B) in theinput voltage V+ of the non-inverting input terminal (+) and the inputvoltage V− of the inverting input terminal (−) of the comparator 623shown in FIG. 6. That is, in the low power supply voltage range in whichthe power supply voltage is low, the input voltage V+ is greater thanthe input voltage V−. But when the power supply voltage is in the rangeof normal values, the input voltage V− is greater than the input voltageV+.

As explained above, this is because the rise in the output voltage ofthe reference voltage generation circuit 621 is set to be early comparedwith the rise in the output voltage of the voltage detection circuit622, and because, on the other hand, after the voltage rise, the outputvoltage of the voltage detection circuit 622 is set to be high comparedwith the output voltage of the reference voltage generation circuit 621.

Consequently, as shown in FIG. 7(C), the output CMPout of the comparator623 goes to H level (high level) in the low power supply voltage rangein which the power supply voltage is low, and goes to L level (lowlevel) when the power supply voltage is in the range of normal values.Very low voltage levels in FIG. 7(C) are induced by unstable rise andfall of power supply.

The output of the comparator 623 is logically inverted by the inverter6241, and so the inverter 6241 outputs a voltage according to the outputof the comparator 623. Consequently the output voltage OUTB of the lowvoltage detection circuit 62 shown in FIG. 6 goes to L level in the lowpower supply voltage range when the power supply voltage is low, andoges to H level when the power supply voltage is in the range of normalvalues, as shown in FIG. 7(D).

This output voltage OUTB is applied to the gate of the MOS transistor M4shown in FIG. 5. As a result, the drain current Id of the MOS transistorM4 becomes the current shown in FIG. 7(E). That is capability isprovided such that the drain current Id of the MOS transistor M4 doesnot flow in the low power supply voltage range in which the power supplyvoltage is low, but flows when the power supply voltage is in the rangeof normal values.

Hence in the low power supply voltage range in which the power supplyvoltage is low, the MOS transistor M4 is turned off. Consequently thereis no outflow of charge which charges the gate of the IGBT 3 to theground line 12 via the MOS transistor M4, and no drop in the gatevoltage of the IGBT 3, and the current Ic flowing in the IGBT 3 is notlimited. On the other hand, in the range of normal values for the powersupply voltage, the MOS transistor M4 is turned on, so that control ofthe collector current Ic of the IGBT 3 by the current control portion 50is maintained.

Next, an example of operation of the low voltage detection circuit 62shown in FIG. 6, which is operation in a case in which the power supplyvoltage momentarily drops, is explained referring to FIG. 5, FIG. 6 andFIG. 8.

As shown in FIG. 8(A), when the voltage Vbin input to the regulatorcircuit 7 momentarily drops, the output voltage Vreg of the regulatorcircuit 7 drops to be approximately equal to the voltage Vbin, but adrop in the internal power supply voltage Vdc is suppressed, as shown inFIG. 8(B).

This is because charge which charges the gate of the IGBT 3 is caused tomove to the power supply line 11 via through diodes D1 and D12 of thecontrol IC 4 e of FIG. 5, and moreover outflow of charge is prevented bythe first protection circuit 30 a.

Here, the gradual drop in the internal power supply voltage Vdc is dueto the fact that charge is supplied to the regulator circuit 7 via theresistor R3 of the first protection circuit 30 a, and simultaneouslycharge flows out due to current consumption of the different circuitswhich use the internal power supply voltage Vdc as a power supply.

This voltage change is accompanied by changes in the input voltage V+ ofthe non-inverting input terminal (+) and in the input voltage V− of theinverting input terminal (−) of the comparator 623 as shown in FIG.8(C). That is, the input voltage V− follows the internal power supplyvoltage Vdc in a state which is made redundant by the amount of theforward-direction voltage of the diode D31, while the input voltage V+is comparatively stable due to the action of the low-pass filter 6212.

Consequently when the voltage Vbin drops momentarily, the input voltageV− of the comparator 623 becomes lower than the input voltage V+ (seeFIG. 8(C)), and so the output CMPout of the comparator 623 momentarilygoes to H level, as shown in FIG. 8(D).

As a result, the output of the inverter 6241 changes from H level to Llevel, so that the charge on the capacitor C10 is momentarily dischargedvia the diode D32. Consequently the output voltage OUTB of the outputcircuit 624 momentarily goes to L level (see FIG. 8(E)), and so the MOStransistor M4 is turned off.

When the voltage Vbin has recovered from a momentary drop, the output ofthe inverter 6241 changes from L level to H level, but due to the filterof the resistor R14 and capacitor C10, the output voltage OUTB risesslowly. By setting the capacitance C10 so as to make this rise timematch the recovery times from the momentary drop of the variouscircuits, the MOS transistor M4 is made to conduct when the variouscircuits are in a normal operating state.

Hence when the voltage Vbin drops momentarily, due to the outflow ofcharge which charges the gate of the IGBT 3 to the ground line 12 viathe MOS transistor M4, the gate voltage of the IGBT 3 drops, and thecurrent Ic flowing in the IGBT 3 is not limited.

Next, waveforms of different portions when the power supply voltage isat a low power supply voltage in the second embodiment are explained,referring to FIG. 9.

In FIG. 9, solid lines are examples of waveforms of different portionsin the second embodiment, and dashed lines are examples of waveforms inan example of the prior art. Further, left-side halves are for a case inwhich the power supply voltage is in the steady state, and right-sidehalves are for a case of a low power supply voltage.

FIG. 9(A) shows the power supply voltage (external power supply voltage)Vbin input to the regulator circuit 7 of FIG. 5. FIG. 9(B) shows acontrol signal Sin (not shown) input to the control IC 4 e of FIG. 5,corresponding to the control signal Sin input to the control IC 4 cshown in FIG. 1. Based on this control signal Sin, switch signals Swpand Swn are generated which turn on and off the MOS transistors M1 andM3 shown in FIG. 5.

FIG. 9(C) shows the output voltage OUTB (solid line) of the low voltagedetection circuit 62 of FIG. 6 and the output voltage OUTB (dashed line)of the low voltage detection circuit 8 of FIG. 15. FIG. 9(D) shows theoutput voltage Vg (gate voltage of the IGBT 3) of the control IC 4 e ofFIG. 5 and the output voltage Vg of the control IC 4 b of FIG. 14. FIG.9(E) shows the collector current Ic of the IGBT 3. FIG. 9(F) shows thevoltage Vref input to the non-inverting amplification terminal of theop-amp 6 of FIG. 5.

On the left side and the right side of FIG. 9(D), the levels of the gatevoltage Vg of the IGBT 3 are different, and on the left side and theright side of FIG. 9(E), the levels of the collector current Ic of theIGBT 3 are different; this is because the drop in the reference voltageVref in FIG. 9(F) causes the op-amp 6 of FIG. 5 to perform currentlimiting operation.

In operation at a low power supply voltage as described above, theoutput voltage OUTB of the low voltage detection circuit 8 shown in FIG.15 goes to H level (see FIG. 16(D)). Consequently, in the example of theprior art, the gate voltage Vg of the IGBT 3 drops, and the collectorcurrent Ic is limited (see the dashed lines of FIGS. 9(C) to 9(F)).

However, in operation at a low power supply voltage such as thatdescribed above in the second embodiment, the output voltage OUTB of thelow voltage detection circuit 62 goes to L level (see FIG. 7(D)).Consequently, in the second embodiment, there is no drop in the gatevoltage Vg of the IGBT 3, and the collector current Ic is not limited(see the solid lines in FIGS. 9(C) to 9(F)).

Next, examples of waveforms of different portions in a case in which thepower supply voltage drops momentarily in the second embodiment areexplained, referring to FIG. 10.

Suppose that the battery voltage Vbatt (see FIG. 1) is in a ripplestate, as shown in FIG. 10(A). In this case, due to resonance of theresonance circuit formed by the wiring inductances L1 and L2 and thecapacitor C1 shown in FIG. 1, a momentary drop occurs in the voltageacross points A-B, as shown in FIG. 10(B). Accompanying this, thevoltage across points C-B is as shown in FIG. 10(C), and changessubstantially similarly to the voltage change across points A-B.

When momentary drops in the voltage across points A-B are repeated as inFIG. 10(B), the low voltage detection circuit 62 repeatedly detects thedrops at the times of each of the momentary drops. Consequently theoutput voltage OUTB of the low voltage detection circuit 62 repeats awaveform in which at each momentary drop, the voltage, after going to Llevel, recovers to H level, as shown in FIG. 10(E).

Consequently, outflow of charge accumulated on the gate of the IGBT 3 isprevented, and drops in the gate voltage Vg of the IGBT 3 are prevented(see FIG. 10(F)), so that sharp declines in the collector current Ic ofthe IGBT 3 can be prevented (see FIG. 10(G)).

Advantageous Results of the Second Embodiment

As explained above, in this second embodiment a first protection circuit30 a and second protection circuit 60 a are provided, as shown in FIG.5. Hence in a state in which a bypass capacitor parallel to the controlIC 4 e is omitted, even when a momentary drop occurs in the power supplyvoltage input to the control IC 4 e, fluctuations in the gate voltage Vgof the IGBT 3 can be suppressed to the extent possible, and adequatedriving of the gate of the IGBT 3 can be secured.

Further, in this second embodiment, when the second protection circuit60 a detects not only a momentary voltage drop in the voltage Vbin ofthe external power supply but also a low power supply voltage state inwhich the power supply voltage is lower than normal, the connectionbetween the current control portion 50 and the ground line 12 isinterrupted. Hence by means of the second embodiment, during operationwhen the power supply voltage is low, drops in the gate voltage Vg ofthe IGBT 3 do not occur, and the IGBT 3 can be driven without limitingthe collector current Ic.

INDUSTRIAL APPLICABILITY

This invention can be applied to driving control of the main switchingdevices of power conversion apparatuses, such as inverters, converters,and similar.

1. A gate driving device which drives a gate of an active element with ahigh gate capacitance, the device comprising: a first switch portionconnected between a first power supply line and a gate of the activeelement and which turns on the active element; a second switch portionconnected between the gate of the active element and a second powersupply line supplied with a lower potential than the first power supplyline and which turns off the active element; a current control portionconnected in parallel with the second switch portion and which controlsoutflow of charge on the gate of the active element to the second powersupply line such that a current that flows in the active element isconstant; a first protection circuit connected between the first switchportion and the gate of the active element and which suppresses outflowof charge on the gate of the active element to the first power supplyline; and a second protection circuit connected between the currentcontrol portion and the second power supply line and which detects aprescribed fluctuation in an applied voltage between the first powersupply line and the second power supply line, and when the fluctuationis detected, interrupts connection between the current control portionand the second power supply line.
 2. The gate driving device accordingto claim 1, wherein the first switch portion comprises a firsttransistor, and the first transistor operates as a constant currentsource when the active element is turned on, and halts operation as theconstant current source when the active element is turned off.
 3. Thegate driving device according to claim 2, wherein the first switchportion further comprises: a second transistor which forms a currentmirror circuit with the first transistor; and a third transistor whichis connected in series with the second transistor and which turns on andoff according to an on or off state of the active element.
 4. The gatedriving device according to claim 1, wherein the second protectioncircuit comprises: a diode which prevents outflow of charge on the gateof the active element to the first power supply line; and a resistorconnected in parallel with the diode and which forms a low-pass filterwith the gate capacitance of the active element.
 5. A gate drivingdevice which drives a gate of an active element with a high gatecapacitance, the device comprising: a first switch portion connectedbetween a first power supply line and the gate of the active element andwhich turns on the active element; a second switch portion connectedbetween the gate of the active element and a second power supply linesupplied with a lower potential than the first power supply line andwhich turns off the active element; a current control portion connectedin parallel with the second switch portion and which controls theoutflow of charge on the gate of the active element to the second powersupply line such that a current that flows in the active element isconstant; a first protection circuit connected between an external powersupply and the first power supply line and which suppresses outflow ofcharge on the gate of the active element to the external power supply;and a second protection circuit connected between the current controlportion and the second power supply line and which, either upondetecting a momentary drop in an external power supply voltage or upondetecting that the external power supply voltage is in a low powersupply voltage state, interrupts connection between the current controlportion and the second power supply line.
 6. The gate driving deviceaccording to claim 5, wherein the second protection circuit comprises: afirst detection circuit which receives as first detection circuit inputvoltage a first protection circuit output voltage and smoothes and thenoutputs the first detection circuit input voltage as a first detectioncircuit output voltage; a second detection circuit which receives asinputs a second detection circuit input voltage and the first protectioncircuit output voltage and generates and then outputs a second detectioncircuit output voltage according to the inputs; a comparator whichcompares the first detection circuit output voltage and the seconddetection circuit output voltage and outputs a comparator output signalaccording to the comparison result; and an output circuit whichgenerates and outputs a prescribed signal based on the comparator outputsignal, and the first detection circuit output voltage is set such thata rise thereof is early compared with a rise of the second detectioncircuit output voltage, and after the rise of the first detectioncircuit output voltage, the first detection circuit output voltage isset to become low compared with the second detection circuit outputvoltage, and a fall amount of the first detection circuit output voltageis set to be small compared with a fall amount of the second detectioncircuit output voltage when the external power supply voltagemomentarily falls.
 7. The gate driving device according to claim 6,wherein the first detection circuit comprises: a first voltage divisioncircuit which divides the first protection circuit output voltage byusing a resistor and a transistor and outputs a first divided voltage;and a low-pass filter which smoothes the first divided voltage.
 8. Thegate driving device according to claim 6, wherein the second detectioncircuit comprises a second voltage division circuit which receives asinputs the first protection circuit output voltage and the firstprotection circuit input voltage, and divides and then outputs theinputs by using a series circuit comprising a first diode, a firstresistor, and a second resistor, and a second diode is connected to theseries connection portion of the first diode and the first resistor. 9.The gate driving device according to claim 6, wherein the secondprotection circuit further comprises a clamp circuit connected betweenoutput terminals of the output circuit.
 10. A gate driving device whichdrives a gate of an active element with a high input capacitance, thedevice comprising: at least one switch portion connected between a powersupply line and a control terminal of the active element and whichswitches the active element; a current control portion connected inparallel with the at least one switch portion and which controls currentflow between the active element and the power supply line to beconstant; and a protection circuit connected between the current controlportion and the power supply line and which detects a prescribed voltagefluctuation, and when the voltage fluctuation is detected, interruptsconnection between the current control portion and the power supplyline.
 11. The gate driving device according to claim 10, wherein theactive element is an insulated gate bipolar transistor (IGBT); the atleast one switch portion includes a first switch portion that turns onthe IGBT, and a second switch portion that turns off the IGBT; the acurrent control portion controls the outflow of charge on the gate ofthe IGBT to a ground line such that a current flowing in the IGBT isconstant; and the protection circuit includes a first protection circuitthat suppresses outflow of charge on the gate of the IGBT to the powersupply line, and a second protection circuit that detects a prescribedfluctuation in an internal power supply voltage, and when thefluctuation is detected, interrupts the connection between the currentcontrol portion and the ground line.